The present invention relates to a method and circuit for creating a modulated signal with a frequency synthesizer based on a phase-locked loop. A fixed-frequency reference signal and a signal created by down-converting from the output signal of the frequency synthesizer are input to the phase comparator of the loop.
A known method of creating a modulated transmitting frequency is shown in FIG. 1. The transmission frequency f.sub.TX is generated in a closed phase-locked loop by controlling a voltage-controlled oscillator 109 (VCO) with a control voltage V.sub.CMOD which is formed from the loop's internal control signal V.sub.C and an external modulation signal MOD, which are summed in a summing circuit 110. As is characteristic of a phase-locked loop, the control voltage V.sub.C is formed with a phase comparator, which is part of the phase-locked circuit 107 (PLL), and a loop filter 108 (LPF) by creating a signal that is proportional to the phase difference between the output signal of the VCO 109 and a reference signal preprogrammed for circuit 107, and then filtering the signal in the loop filter 108 (LPF). Only those phase or frequency changes caused by the modulating signal MOD that are faster than the settling time of the loop can be detected in the output signal f.sub.TX of the VCO, which means that the loop has a high pass type frequency response to the modulating signal MOD.
FIG. 1 also shows a conventional heterodyne receiver in which the receiver signal f.sub.RX is mixed with a local frequency f.sub.1 in a mixer 3 and filtered with a passband filter 4 to form an intermediate frequency f.sub.IF. A local frequency f.sub.1 is generated in an oscillator 5, which may be a phase-locked frequency synthesizer or a crystal oscillator, for example. The antenna 1 of the transceiver, a duplex filter 2 and the power amplifier 106 of the transmitter are also shown in the figure for clarity.
The frequency adjustment range of voltage-controlled oscillators used in mobile phones is about 30 MHz (e.g., when the carrier frequencies are 890-915 Mhz) and the control voltage range is less than 5 V, whereupon a one kilohertz deviation in the output frequency of the VCO corresponds to a 200 .mu.V variation in control voltage and, as a consequence, noise in modulation signal MOD easily cause inaccuracy in output signal f.sub.TX. Furthermore, the modulation coefficient defined as the ratio between the frequency change of VCO 109 and the corresponding change in control voltage V.sub.CMOD is not constant but varies as VCO output frequency f.sub.TX changes. As a result, the modulation coefficient changes easily from device to device and in proportion to frequency or temperature. The solution shown in FIG. 1 is advantageous from the point of view of the transmitter's energy efficiency, because the output power of the VCO 109 is applied almost directly to the transmission signal f.sub.TX. The solution is simple but not suitable for DC modulation or digital phase modulation due to the high pass nature of its frequency response and its sensitivity to noise.
The problems of the solution shown in FIG. 1 can be partially eliminated with the solution shown in FIG. 2, in which the transmission frequency f.sub.TX is formed by mixing a common local frequency f.sub.1 of the transceiver unit and a modulated offset frequency f.sub.OFF generated in an oscillator 209. The modulation coefficient can be realized according to practical requirements, whereupon, for example, a frequency deviation of about 1 KHz can be created with approximately a one volt variation in the modulating signal MOD. Either a frequency f.sub.1 +f.sub.OFF or f.sub.1 -f.sub.OFF at the output of mixer 208 is selected with passband filter 207 as the transmission frequency f.sub.TX. The signal is attenuated approximately 10 dB in the mixer 208 and the filter 207, thus reducing the overall efficiency of the circuit when compared to the solution in FIG. 1. The signal outputted by the filter 207 is applied by amplifier 206 to the duplex filter 2. If the oscillator 209 is a crystal oscillator, for example, this construction can be used to create DC modulation. The frequencies of crystal oscillators that can be modulated are typically below 50 MHz, limiting the applicability of the solution shown in FIG. 2. The solution has further problems because it is not accurate enough for phase modulation.
The I/Q modulator 310 shown in FIG. 3 is suitable for creating controlled phase modulation. Unmodulated, constant-amplitude, transmitting frequency signal f.sub.C and signal f.sub.Cp/2 having the same amplitude and a 90-degree phase shift as compared to signal f.sub.C are multiplied by modulating signal components I and Q in multipliers 308 and 309, respectively. The resulting signals are added together in a summing circuit 307, the output of which is the modulated transmitting frequency f.sub.TX to be applied by amplifier 306 to the duplex filter 2. The problem with this circuit is that the I/Q modulator 310 is difficult to implement, especially at high frequencies, because multipliers 308 and 309 must have good linearity. Multipliers 308, 309 and summing circuit 307 are usually implemented as active circuits, such as well-known Gilbert cell multipliers, whose current consumption may be tens of milliamperes. Another problem with this circuit is that it is difficult to prevent the unmodulated signal f.sub.C of transmission frequency from leaking through the multiplier into the transmission. For these reasons, instead of I/Q modulation at transmitting frequency, I/Q modulation at offset oscillator frequency f.sub.T, as shown in FIG. 4, is often used. A modulated offset oscillator frequency f.sub.OFF is up-converted with the help of the local frequency f.sub.1 to transmission frequency f.sub.TX which is applied by amplifier 406 to the duplex filter 2. In this circuit the modulator circuit 410 is operated at a much lower frequency than the transmission frequency, such as 90 MHz, while the transmission frequency f.sub.TX is about 900 MHz. The transmission frequency f.sub.TX is selected with filter 412 from one of the output frequencies of mixer 411, f.sub.1 +f.sub.OFF or f.sub.1 -f.sub.OFF, that are mixed from frequencies f.sub.1 and f.sub.OFF. The filter also attenuates leakage of frequencies f.sub.1 and f.sub.OFF to the transmitter. The current consumption of the structure shown in FIGS. 3 and 4 is high, mainly due to the multipliers and summing circuits 307 and 407 of I/Q modulators 310, 410, and the loss of signal energy in filter 412 of FIG. 4.